Dual mode power supply controller with charge balance multipliers and charge balance multiplier circuits

ABSTRACT

A circuit for generating an output current includes a control signal generating circuit that is configured to generate a control signal. The control signal is a function of a level of an analog input voltage signal, and a level of the output current is a function of a level of an analog input current signal and the level of the analog input voltage signal.

RELATED APPLICATION

The present application is a continuation-in-part of U.S. patentapplication Ser. No. 13/664,979, filed on Oct. 31, 2012 , the disclosureof which is hereby incorporated by reference herein as if set forth inits entirety.

TECHNICAL FIELD

The present disclosure relates to power converter circuits, and moreparticularly to power converter circuits that generate output power.

BACKGROUND

Power converters, or power supplies, may be used in electronicapplications to convert an input voltage to a desired output voltage topower one or more electronic devices. Some power supplies may beclassified as either a linear power supplies or a switched-mode powersupply (SMPS).

Switched-mode power supplies may be configured to operate moreefficiently than linear power supplies. A switched-mode power supply mayinclude a switch that, when switching on and off, stores energy in aninductor and discharges the stored energy to an output of the switchedmode power supply. The switch may be controlled by a controller, whichoutputs switching signals to turn the switch on and off.

SUMMARY

A circuit for generating an output current includes a control signalgenerating circuit that is configured to generate a control signal. Thecontrol signal is a function of a level of an analog input voltagesignal, and a level of the output current is a function of a level of ananalog input current signal and the level of the analog input voltagesignal.

The circuit may further include a switched current source that iscontrolled by the control signal, and a balance capacitor coupled to theswitched current source. A duty cycle of the control signal may beproportional to a level of the analog input voltage signal, and a levelof the output current may be proportional to a product of a level of theanalog input current signal and the level of the analog input voltagesignal.

The circuit may further include an output current mirror coupled to thebalance capacitor. The

switched current source may receive the analog input current signal asan input, and

the balance capacitor may be charged by a current output by the switchedcurrent source and discharged through the output current mirror.

The circuit may further include a resistor between the balance capacitorand the output current mirror.

The control signal generating circuit may include a comparatorconfigured to receive the analog input voltage signal and a ramp voltageand to generate the control signal in response to the analog inputvoltage signal and the ramp voltage, and a ramp voltage generatingcircuit coupled to the comparator and configured to generate the rampvoltage.

The ramp voltage generating circuit may include a current sourceconfigured to generate a ramping current, a ramping capacitor coupled tothe current source and configured to be charged by the ramping current,a transistor switch configured to discharge the ramping capacitor inresponse to a discharge signal, and a hysteretic comparator configuredto compare the ramping voltage with a reference voltage and to generatethe discharge signal in response to the comparison of the rampingvoltage with the reference voltage.

The current mirror may include a first transistor having a gate terminaland a drain terminal, a second transistor having a gate terminal coupledto the gate terminal of the first transistor and a drain terminalcoupled to the drain terminal of the first transistor, and a switchtransistor coupled between the drain terminals of the first and secondtransistors and the gate terminals of the first and second transistors.The switch terminal has a gate terminal coupled to an output of thecomparator and is configured to receive the control signal.

The circuit may further include an input current conditioning circuitincluding a first current mirror and a second current mirror coupled tothe first current mirror. The first current mirror may be configured tosupply an input current signal as the analog input current signal whenthe analog input current signal is above a threshold level, and thesecond current mirror may be configured to supply a reference currentsignal as the analog input current signal when the analog input currentsignal is below the threshold level.

The multiplier circuit may be configured to be switched between a firstmode in which the input current signal is nonzero and a second mode inwhich the input current signal is zero.

The circuit may further include a clamping diode coupled to the balancecapacitor.

The output current may be given as I_(PK)=(V_(COMP)*I_(CH))V_(LIMIT),where I_(PK) is the output current, V_(COMP) is the analog input voltagesignal, I_(CH) is the analog input current signal, and V_(LIMIT) is areference voltage.

A charge that is stored in the balance capacitor may be given as(I_(CH)−I_(PK))DTs, where D is a duty cycle of the control signal and Tsis a period of the control signal, and wherein a charge that isdischarged from the balance capacitor is given as I_(PK)(1−D)Ts.

The switched current source may receive a reference current as an input,and the balance capacitor may be charged by the analog input currentsignal and discharged by the switched current source.

The circuit may further include a hysteretic inverter having an inputcoupled to the balance capacitor, and a switch coupled to an output ofthe hysteretic inverter and configured to control the switched currentsource.

The circuit may further include an output inverter having an inputcoupled to the output of the hysteretic inverter and configured togenerate an output signal having an amplitude that is proportional tothe analog input voltage signal.

The circuit may further include a filter configured to filter the outputsignal of the output inverter, and an amplifier configured to amplifythe filtered output signal of the output inverter.

The output current signal may be given by I_(PK)=I_(CH)*V_(COMP)*K,wherein I_(PK) is the output current, I_(CH) is the analog input currentsignal, V_(COMP) is the analog input voltage signal, and K is aconstant.

A charge that is stored in the balance capacitor may be given asI_(CH)(1−D)Ts, where D is a duty cycle of the control signal and Ts is aperiod of the control signal, and wherein a charge that is dischargedfrom the balance capacitor is given as (I_(FS)−I_(CH))DTs, whereinI_(FS) is the reference current.

A power conversion circuit according to some embodiments includes avoltage boost circuit including a boost inductor, the voltage boostcircuit being configured to generate an output voltage in response to aninput voltage, and a boost controller configured to control operation ofthe voltage boost circuit.

The boost controller is configured to generate an error signal mymultiplying a current signal and a voltage signal, and the powerconversion circuit further comprises a multiplier circuit formultiplying the current signal by the voltage signal. The multipliercircuit includes a switched current source that is controlled by acontrol signal, a control signal generating circuit that is configuredto generate the control signal, wherein a duty cycle of the controlsignal is proportional to a level of the voltage signal, and a balancecapacitor coupled to the switched current source. A level of the outputcurrent is proportional to a product of a level of the current signaland the level of the voltage signal.

The power conversion circuit may further include an output currentmirror coupled to the balance capacitor. The switched current source mayreceive the analog input current signal as an input, and the balancecapacitor may be charged by a current output by the switched currentsource and is discharged through the output current mirror.

The switched current source may receive a reference current as an input,and the balance capacitor may be charged by the analog input currentsignal and discharged by the switched current source.

It is noted that aspects of the inventive concepts described withrespect to one embodiment may be incorporated in a different embodimentsalthough not specifically described relative thereto. That is, allembodiments and/or features of any embodiments can be combined in anyway and/or combination. These and other objects and/or aspects of thepresent inventive concepts are explained in detail in the specificationset forth below.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings are included to provide a furtherunderstanding of the disclosure and are incorporated in and constitute apart of this application. In the drawings:

FIG. 1 is a block diagram of a power converter circuit according to someembodiments.

FIG. 2 is a circuit diagram of a power converter circuit according tosome embodiments.

FIG. 3 is a block diagram of a control circuit for a power convertercircuit according to some embodiments.

FIG. 4 is a conceptual timing diagram illustrating the timing of variousvoltage and current signals within a power converter circuit including acontrol circuit as shown in FIG. 3.

FIG. 5 is a block diagram of a control circuit for a power convertercircuit according to further embodiments.

FIG. 6 is a conceptual timing diagram illustrating the timing of variousvoltage and current signals within a power converter circuit including acontrol circuit as shown in FIG. 3.

FIG. 7 is a circuit diagram of a power converter circuit according tofurther embodiments.

FIG. 8A is a circuit diagram of a multiplier and limiter circuit for apower converter circuit as shown in FIG. 7 and configured to operate inhysteretic mode.

FIG. 8B is a conceptual timing diagram illustrating the timing ofvarious voltage and current signals within a power converter circuit asshown in FIG. 7 and configured to operate in hysteretic mode.

FIG. 9A is a circuit diagram of a multiplier and limiter circuit for apower converter circuit as shown in FIG. 7 and configured to operate incritical current mode.

FIG. 9B is a conceptual timing diagram illustrating the timing ofvarious voltage and current signals within a power converter circuit asshown in FIG. 7 and configured to operate in critical current mode.

FIGS. 10-16 are flowcharts illustrating operations of circuits/methodsaccording to some embodiments.

FIG. 17 is a schematic diagram of a charge balance multiplier inaccordance with some embodiments.

FIGS. 18A and 18B are graphs that illustrate operation of the chargebalance multiplier of FIG. 17.

FIG. 19 is a graph illustrating the relationship of the duty cycle ofthe charging current and the ramping voltage of the circuit of FIG. 17.

FIG. 20 is a schematic diagram of a charge balance multiplier inaccordance with further embodiments.

DETAILED DESCRIPTION

Embodiments of the present inventive concepts now will be described morefully hereinafter with reference to the accompanying drawings. Theinventive concepts may, however, be embodied in many different forms andshould not be construed as limited to the embodiments set forth herein.Rather, these embodiments are provided so that this disclosure will bethorough and complete, and will fully convey the scope of the inventiveconcepts to those skilled in the art. Like numbers refer to likeelements throughout.

It will be understood that, although the terms first, second, etc. maybe used herein to describe various elements, these elements should notbe limited by these terms. These terms are only used to distinguish oneelement from another. For example, a first element could be termed asecond element, and, similarly, a second element could be termed a firstelement, without departing from the scope of the present inventiveconcepts. As used herein, the term “and/or” includes any and allcombinations of one or more of the associated listed items.

The terminology used herein is for the purpose of describing particularembodiments only and is not intended to be limiting. As used herein, thesingular forms “a”, “an” and “the” are intended to include the pluralforms as well, unless the context clearly indicates otherwise. It willbe further understood that the terms “comprises,” “comprising,”“includes” and/or “including” when used herein, specify the presence ofstated features, integers, steps, operations, elements, and/orcomponents, but do not preclude the presence or addition of one or moreother features, integers, steps, operations, elements, components,and/or groups thereof.

Unless otherwise defined, all terms (including technical and scientificterms) used herein have the same meaning as commonly understood by oneof ordinary skill in the art to which this disclosure belongs. It willbe further understood that terms used herein should be interpreted ashaving a meaning that is consistent with their meaning in the context ofthis specification and the relevant art and will not be interpreted inan idealized or overly formal sense unless expressly so defined herein.

FIG. 1 illustrates a power converter 10 according to some embodiments.The power converter 10 receives an AC input voltage V_(AC) (which may,for example be a 110, 220 or 240 volt AC line voltage) and converts theinput voltage to a boosted DC signal V_(BOOST) that is used to drive aload 40. The power converter 10 includes a rectification and filteringcircuit 20 that generates a rectified and filtered voltage V_(RECT) inresponse to the input voltage, and a voltage boost circuit 30 thatgenerates the boosted DC signal V_(BOOST) in response to the rectifiedand filtered voltage V_(RECT).

Some embodiments provide a voltage boost circuit that regulates a levelof current supplied to the load 40. Regulating the load current may beparticularly important when driving solid state lighting devices,because the color and/or intensity of light emitted by LEDs may beaffected by the level of current flowing through the devices. Variationsin drive current may therefore result in undesirable variations in thecolor and/or intensity of the light output by the apparatus.

FIG. 2 illustrates a power converter 10 including an integrated circuit(IC) controller 100 that regulates load current supplied to the LED load40. The power converter 10 is powered by a regulated AC input 22 shownas a sine wave voltage generator 22. The input AC signal is rectified bya diode bridge D1 and filtered by a capacitor C1 in the rectificationand filtering circuit 20. The output of the rectification and filteringcircuit 20 is a rectified sine wave V_(RECT).

The boost converter includes a boost inductor, L1, an output capacitorC2, a diode D2 and a cascade switch including first and second switchesQ1 and Q2. A resistor R1 and a Zener diode D4 provide a bias supply forthe first switch Q1.

Because the gate of the first switch Q1 is biased by the Zener diode D4,the conductivity of the first and second switches Q1 and Q2 iscontrolled by a pulse width modulation (PWM) signal applied to the gateof the switch Q2. When the switches Q1 and Q2 are ON, the boost inductorL1 is coupled to ground, causing current through the boost inductor L1to increase, which stores energy in the boost inductor L1. When theswitches are turned off, energy stored in the boost inductor L1 isdischarged through the diode D2 to charge the output capacitor C2. Byregulating the frequency and/or duration of PWM pulses applied to thegate of the second switch Q2, the voltage level on the output capacitorC2 can be controlled.

A diode D3 and a capacitor C3 work in combination with the cascadeswitch and a bias regulator 52 to generate a bias signal VCC that may beused, for example, to power the IC controller 100. A suitable biasregulator 52 is described in detail in co-pending and commonly assignedU.S. application Ser. No. 13/664,895 filed concurrently herewith, thedisclosure of which is incorporated herein by reference.

The load current is regulated by monitoring the load current and thecurrent in the boost inductor L1. The resistors R2 and R3 are used bythe controller 100 to monitor the return inductor current via a currentsense pin CS of the controller 100. The resistor R5 is used by thecontroller 100 to monitor the load current via a feedback pin FB of thecontroller 100. In particular, an error amplifier 70 generates an errorsignal that represents the difference between the actual load currentand a target load current value. A pulse width modulator 60 generatesthe signal PWM that controls the conductivity of the cascade switch inresponse to a level of the error signal.

Embodiments of the present invention are based on the realization thatin some cases it may be desirable to regulate the current supplied tothe load 40 instead of regulating the voltage applied to the load 40.

As discussed in more detail below, the controller 100 may be configuredto operate in either a hysteretic current mode or a critical currentmode. In the hysteretic current mode, the inductor current is controlledto operate within a predetermined range based on the level of the errorsignal. In the critical current mode, the error signal is allowed tofluctuate with the level of the rectified input voltage. The inductorcurrent is thereby controlled to have a peak value that is generallyproportional to the level of the rectified input voltage V_(RECT).

As shown in FIG. 2, the controller 100 includes a current sensingcircuit 56 that senses a level of the inductor current. The currentsensing circuit 56 may also generate a FEED FORWARD signal thatrepresents the level of the rectified input voltage V_(RECT) when thecontroller is configured for operation in critical current mode, and/ora hysteretic current mode signal HYS when the controller is configuredfor operation in hysteretic current mode.

A resistor R4 is connected to a selection pin HYS-FF of the controller100 and to a switch S1. The function of the resistor R4 depends upon onthe setting of switch S1. That is, switch S1 may connect the resistor R4from pin HYS-FF to ground or to the V_(RECT) signal output by therectification and filtering circuit 20.

Connecting the resistor R4 to ground places the controller in hystereticcurrent mode and defines a hysteresis window within which the inductorcurrent can fluctuate. In contrast, connecting the resistor R4 from pinHYS-FF to V_(RECT) places the controller 100 in the critical currentmode and provides a V_(RECT) feed forward signal to the HYS-FF pin ofthe controller 100.

When the HYS-FF pin is connected to ground through switch S1, thecurrent sensing circuit 56 generates a HYS signal having a HIGH level,which configures the IC for operation in the hysteretic current mode.When the HYS-FF pin is connected to V_(RECT) through switch S1, thecurrent sensing circuit 56 generates a HYS signal having a LOW level,which configures the IC for operation in the critical current mode.

It will be appreciated that the physical switch S1 is optional. That is,the resistor R4 may be connected to either ground or V_(RECT) by hardwiring the connection.

According to some embodiments, the elements of the controller 100 may beformed on a single integrated circuit chip. The single chip controller100 may include the second switch Q2 of the cascade switch, the biasregulator 52, the PWM modulator 60, the current sensing circuit 56, aprotection circuit 54, a limiter 62, the error amplifier 70, a limitingdiode D5, and a multiplier 64.

The PWM modulator 60 compares the inductor current with an ERROR signal(which in the hysteretic mode is the COMP signal output by the erroramplifier 70) and responsively generates the PWM signal that drives thesecond switch Q2.

The protection circuit 54 generates an INHIBIT signal that stopsoperation of the PWM modulator 60 in response to one or more statusindications, such as a low bias power indication, an over temperatureindication, etc.

The error amplifier 70 compares the load current sensed at the senseresistor R5 with a reference voltage V_(REF) and outputs a comparisonsignal COMP that is proportional to the difference between them. Thecontroller 100 attempts to control the inductor current so that theactual output current stays close to a target output current defined byV_(REF)/R5. The level of the COMP signal is limited by the Zener diodeD5, which effectively provides current limiting of the current throughinductor L1.

In the critical current mode, (when HYS=LOW) the output signal COMP ofthe error amplifier 70 is multiplied by the FEED FORWARD signal outputby the current sensing circuit 56 in a multiplier 64. This causes thepeak inductor current to follow V_(RECT). The output of the multiplier64 is applied to a limiter 62, which may limit both the upper and lowerbounds of the ERROR signal. By limiting the range of the ERROR signal,the limiter 62 limits the inductor current.

In the hysteretic current mode (when HYS=HIGH), the control switch S2causes the COMP signal to bypass the multiplier 64 in response to theHYS signal. This causes the COMP signal to be applied directly to thePWM modulator 60 as the ERROR signal. In the hysteretic current mode,the peak inductor current is therefore proportional to the level of theCOMP signal, except as limited by the diode D5.

FIG. 3 is a circuit diagram that illustrates a current sensing circuit56, a PWM modulator 60, and an error amplifier 70 according to someembodiments. In the embodiments illustrated in FIG. 3, the resistor R4is connected from the HYS-FF pin to ground, which places the controller100 in the hysteretic current mode. The limiter 62 is omitted forclarity as it is bypassed in the hysteretic mode.

In the embodiments of FIG. 3, the current sensing circuit 56 includesamplifiers 74, 76, an NMOS transistor Q3 at the output of the amplifier74, an NMOS transistor Q4 at the output of the amplifier 76, and currentmirrors 72, 82 coupled to the drains of transistors Q3, Q4,respectively. The inverting input of the amplifier 74 is coupled to theCS pin along with the source of the NMOS transistor Q3. Thenon-inverting input of the amplifier 74 is coupled to ground.

The inverting input of the amplifier 76 is coupled to the HYS-FF pinalong with the source of the NMOS transistor Q4. The non-inverting inputof the amplifier 76 is coupled to a reference voltage V_(BG).

A PMOS transistor Q5 has a source coupled to the HYS-FF pin and a draincoupled to a feed forward sense resistor R6. The gate of the PMOStransistor Q5 is coupled to the output of the amplifier 76.

The amplifier 76 attempts to hold the voltage at the gates of thetransistors Q4 and Q5 at a voltage that is equal to the sum of V_(BG)and the threshold voltage of the NMOS transistor Q4. Meanwhile, thevoltage at the HYS-FF pin is held at V_(BG). This causes the NMOStransistor Q4 to be ‘on’ and the PMOS transistor Q5 to be ‘off,’ andcauses a current I_(HYS) to flow through the resistor R4 at the HYS-FFpin.

The current mirror 82 outputs a current that is approximately equal tothe current I_(HYS) flowing through the HYS-FF pin.

The return inductor current flowing through the resistor R2 causes thevoltage at node 68 to be negative. Meanwhile, the amplifier 74 isreferenced to ground, which causes the amplifier to hold the gate of thetransistor Q3 at a voltage that is greater than zero, which turns on thetransistor Q3 and causes a current I_(CS) to flow out the CS pin andthrough the resistor R3. The current I_(CS) is proportional to thecurrent flowing through the inductor L1.

The current mirror 72 includes a first output line 73 and a secondoutput line 75. A copy of the current I_(CS) flowing through the currentsense pin CS is output by the current mirror 72 on the first output line73 and the second output line 75. The current on the first output lineis sensed by a first current sense resistor R7. The output I_(HYS) ofthe current mirror 82 is combined with the current I_(CS) on the secondoutput line 75 at a combination node 77, and the combined currentI_(CS)+I_(HYS) is sensed by a second sense resistor R8. The first senseresistor R7 and the second sense resistor R8 may have the sameresistance value R. The feed forward sense resistor R6 may also have thesame resistance value R.

The voltage sensed by the first sense resistor R7 is applied to thenon-inverting input of a comparator 94, while the voltage sensed by thesecond sense resistor R8 is applied to the inverting input of acomparator 98. The comparator 94 generates a RESET signal to reset theoutput of a latch 95 to LOW, while the comparator 98 generates a SETsignal to set the output of the latch 95 HIGH. The RESET signal iscombined with the INHIBIT signal in an OR-gate 96. Accordingly, the PWMsignal may be reset in response to either a RESET signal generated bythe comparator 94 or the INHIBIT signal generated by the protectioncircuit 54 (FIG. 2).

The output of the latch 95 is provided as the PWM signal to the secondtransistor switch Q2.

The current sensing circuit 56 further includes a MODE comparator 78that outputs the HYS signal in response to the connection of the HYS-FFpin to ground. That is, when the hysteretic mode is selected, thecomparator 76 causes the voltage at the gate of Q4 (which is connectedto the noninverting input of the comparator 78) to exceed the voltage atthe source of Q4 (which is connected to the inverting input of thecomparator 78). In that case, the comparator 74 outputs a HIGH voltagelevel as the HYS signal.

The error amplifier circuit 70 includes an amplifier 92 having aninverting input coupled to the feedback pin FB, a non-inverting inputcoupled to a reference voltage V_(REF), and an output coupled to alimiting Zener diode D5. When the HYS signal is HIGH, the multiplier isbypassed by the switch S1, and the COMP signal output by the erroramplifier 70 becomes the ERROR signal input to the comparators 94, 98.

The voltage at the HYS-FF pin is held at V_(BG) to forward bias the NMOStransistor Q4. The current flowing in the resistor R4 is thereforeI_(HYS)=V_(BG)/R4. The PMOS transistor Q5 is biased ‘off’, which causesthe feed forward current I_(FF) to be zero. The MODE comparator 78monitors the gate-to-source voltage of both FETs Q4 and Q5. Because theNMOS transistor Q4 is ‘on’ and the PMOS transistor Q5 is ‘off’, theoutput HYS of the MODE comparator 78 is HIGH. The voltage controlledswitches S2 and S3 are set by the MODE signal in the positions shown inFIG. 3. Namely, the switch S2 connects the COMP output of the erroramplifier to the ERROR input of the PWM modulator 60, and the switch S3connects the non-inverting input of the comparator 98 to the ERRORsignal.

In operation, the return inductor current is monitored across resistorR2. The voltage at CS pin is held at ground, such that the current in R3(and the CS pin) is proportional to the inductor current

$\left( {I_{CS} = {I_{L\; 1} \cdot \frac{R\; 2}{R\; 3}}} \right).$The current I_(CS) is output by the current mirror 72 on output line 73,and a current I_(HYS) is added to the current I_(CS), and the combinedcurrent I_(HYS)+I_(CS) is output on line 75. The current I_(CS) issensed at sense resistor R7, while the current I_(HYS)+I_(CS) is sensedat sense resistor R8. Accordingly, when the current I_(Cs)+I_(HYS) fallsto a level such that the voltage sensed at resistor R8 is less than theERROR signal output by the error amplifier 70, the SET signal output bythe comparator 98 transitions to HIGH, causing the PWM signal output bythe latch 95 to transition to HIGH.

Likewise, when the current I_(CS) rises to a level such that the voltagesensed at resistor R7 is greater than the ERROR signal output by theerror amplifier 70, the RESET signal output by the comparator 94transitions to HIGH, causing the PWM signal to transition to LOW.

FIG. 4 is a timing diagram showing the periodic steady-state operationof the circuit of FIG. 3. In particular, FIG. 4 shows relative timing ofthe SET, RESET, PWM signals as well as the current levels of I_(CS) andI_(CS)+I_(HYS). A copy of I_(CS) as scaled by R7 is compared to theERROR signal to reset the PWM latch 95. That is, the PWM signal goes LOWwhen I_(CS)*R7 exceeds the level of the ERROR signal. A copy of I_(HYS)is added to a copy of I_(CS) and the resulting summation(I_(CS)+I_(HYS)) is scaled by R8 for comparison with ERROR to set thePWM latch 95 (PWM goes HIGH).

FIG. 5 is a block diagram similar to FIG. 3 showing the current sensingcircuit 56, the PWM modulator 60, the limiter 62 and the error amplifier70 except that in FIG. 5, the resistor R4 is connected from the HYS-FFpin to V_(RECT), which places the circuit into the critical currentmode. The limiter 62 is illustrated in FIG. 5, and includes V_(LOW) andV_(HIGH) reference voltages and diodes D6 and D7.

In the critical current mode, the voltage at the HYS-FF pin is held atV_(BG) to forward bias the Q5 PMOS gate, causing a feed forward currentI_(FF) to flow in R4. The NMOS transistor Q4 is biased ‘off’ andI_(HYS)=0. The output signal HYS of the MODE comparator is LOW, causingthe voltage controlled switches S2 and S3 to be in the positions shownin FIG. 5. Namely, the switch S2 connects the output of the multiplier64 to the ERROR input of the PWM modulator 60, and the switch S3connects the non-inverting input of the comparator 98 to GND.

In the critical current mode, when the voltage sensed at R8 (equal toI_(CS)*R) drops below zero, the comparator 98 outputs a HIGH voltage,causing the latch 95 to transition to HIGH. When the voltage sensed atR7 (also equal to I_(CS)*R) exceeds the ERROR voltage, the comparator 94outputs a HIGH voltage, causing the latch 95 to transition to LOW. Inthe critical current mode, the ERROR voltage follows the V_(RECT)voltage with a floor at the V_(LOW) voltage level and a ceiling at theV_(HIGH) voltage level due to the limiter 62.

FIG. 6 shows the periodic steady-state operation for the criticalcurrent mode. A copy of I_(CS) output on line 73 and scaled by R iscompared to ERROR to reset the PWM latch (causing the PWM signal to goLOW). Another copy of I_(CS) output on line 75 is scaled by R (due toI_(HYS)=0) and compared to ground to set the PWM latch (PWM goes HIGH).Note that I_(HYS) is zero because of the connection of resistor R4 toV_(RECT). The valley of the inductor ripple current reduces to zero incritical current mode.

In the critical current mode, the FEED FORWARD signal is multiplied bythe output of the error amplifier 92 to generate the ERROR signal.Connecting the resistor R4 to V_(RECT) modulates the ERROR signal andcauses the peak inductor current to follow a rectified sine wave. Thefeed forward current I_(FF) is equal to (V_(RECT)−V_(BG))/R4). Theamplitude of V_(RECT) is large compare to V_(BG) for most of the cycleand V_(BG) can be neglected. The Feed Forward current I_(FF) and theFeed Forward voltage signal (=I_(FF)/R) approximate a rectified sinewave. The integration time constant of the error amplifier is very lowso that the COMP output of the error amplifier 70 can be consideredconstant. The ERROR signal is the product of the Feed Forward voltagesignal and COMP. The current I_(CS), which is proportional to theinductor current, is compared to the ERROR signal to reset the PWMlatch, causing the PWM signal to go LOW. The inductor current peakfollows the rectified voltage wave shape.

The limiter 62 limits the range of the ERROR signal. The upper bound forERROR is V_(HIGH), which sets the maximum input current. The lower boundfor ERROR in the critical current mode is V_(LOW), which sets theminimum peak inductor current during the zero-crossing of the AC input.

A controller 100A according to further embodiments is illustrated inFIG. 7. Like the controller circuit illustrated in FIGS. 2, 3 and 5, thecontroller 100A causes a voltage boost circuit to supply a controlledamount of current to a load 40. However, the controller 100A shown inFIG. 7 differs from the circuits shown in FIGS. 2, 3 and 5 in that thecontroller 100A uses current control signals to generate the PWM signal.

In particular, the controller 100A includes a current sensing block 56Athat is similar to the current sensing block 56 of FIGS. 2, 3 and 5,except that the current sensing block 56A does not include the senseresistors R7 and R8. Furthermore, the voltage comparators 94 and 98 inthe PWM Modulator are eliminated. In the controller 100A, currentcomparators or simple logic gates are used to provide switching signals.It will be appreciated that a positive current flowing into a logic gateinput produces a HIGH voltage at the input of the logic gate, while anegative current at the input of a logic gate results in a LOW voltageat the input of the gate. This aspect of logic gates may be exploited toprovide a circuit that uses current signals rather than voltage signalsto control a PWM signal.

In the hysteretic mode, the current sensing block 56A generates anoutput current equal to ICS on line 73 and an output current equal toI_(CS)+I_(HYS) on line 75. The current sensing block 56A also generatesa mode signal CrCM as an output of the comparator 78 in the mannerdescribed above to generate the HYS signal. In the critical currentmode, the current sensing block also generates a feedforward currentI_(FF) that is proportional to V_(RECT), as described above.

The use of current control signals can be more accurate in someapplications depending upon the specific silicon process used tofabricate the controller. Typical silicon processes used formixed-signal power control ICs rely on matching devices to meet theaccuracy requirements. In the case of the circuit of FIGS. 3 and 5, thePWM modulator is controlled by voltages across sense resistors R7 andR8, which are assumed to have the same resistance. However, the absolutetolerance of these resistors can be over ±30% as a result of temperatureand process variations. (Note, however, that similar resistors on asingle integrated circuit will track each other such that the relativematching accuracy can achieve ±0.1%.) There are options availableincluding thin film processing and trimming techniques that can improvethe absolute tolerance, but these typically add cost. The use of currentcontrol signals can provide an approach that meets the requiredaccuracy.

The controller 100A shown in FIG. 7 further includes an error amplifier70 that functions in a similar manner as described above to produce avoltage signal V_(COMP). A multiplier and limiter circuit 110 generatesan output currents I_(PK) and I_(VAL), as a function of V_(COMP), CrCM,and I_(FF). The value of I_(VAL) is subtracted from the sum of I_(CS)and I_(HYS), and the result is applied to the input of an inverter 112.The value of I_(PK) is subtracted from I_(CS), and the result is appliedto the input of a buffer 114.

Accordingly, when I_(CS)+I_(HYS)−I_(VAL) falls to zero, the input of theinverter 112 is LOW, which causes the output of the inverter 112 to goHIGH, which sets the PWM latch 95. That is, when I_(CS)+I_(HYS) falls toa value that is equal to I_(VAL) or lower, a negative current is drawnfrom the input of the inverter 112, causing its output to transition toHIGH.

Similarly, the latch 95 is reset when the output of the buffer 114transitions to HIGH. When I_(CS)−I_(PK) is positive, (i.e. when I_(CS)exceeds I_(PK)), a positive voltage appears at both the input and outputof the buffer 114. Because the output of the buffer 114 is provided tothe RESET input of the latch 95, this resets the PWM latch 95. I_(PK)and I_(VAL) are therefore similar in function to the ERROR signaldescribed in connection with FIGS. 2-6.

FIG. 8A is a circuit diagram of a multiplier and limiter circuit 110 fora power converter circuit as shown in FIG. 7 and configured to operatein hysteretic mode. FIG. 8B is a conceptual timing diagram illustratingthe timing of various voltage and current signals within a powerconverter circuit as shown in FIG. 7 and configured to operate inhysteretic mode.

As shown in FIG. 8A, the multiplier and limiter circuit 110 includes amultiplier 99 that receives the V_(COMP) signal output by the comparator92 and the I_(FF) signal. In the hysteretic mode, the I_(FF) signal iszero and I_(PK-LOW) is constant, so the multiplier simply applies afixed gain Gm to the V_(COMP) signal. The output is passed to a limiter97 that generates a current I_(PK) in response to the multiplier signal.The I_(PK) current signal is limited at an upper bound of I_(PK-HIGH). Acurrent mirror 98 is connected to the limiter 97 and generates twocurrent signals I_(PK) and I_(VAL) that are equal to the current outputby the limiter 97. The current signals I_(PK) and I_(VAL) are subtractedat nodes 84 and 86 from the I_(CS) and I_(CS)+I_(HYS) signals,respectively.

Referring to FIG. 8B, when the value of I_(CS)+I_(HYS) drops below thereference value of I_(VAL), the current into the input of the inverter112 is negative, which produces a LOW voltage at the input of theinverter 112 and a HIGH voltage at the output of the inverter 112. Thissets the PWM signal to a HIGH level. When the value of I_(CS) risesabove the reference value of I_(PK), the current into the input of thebuffer 114 is positive, which produces a HIGH voltage at the input ofthe buffer 114 and a HIGH voltage at the output of the buffer 114. Thisresets the PWM signal to a LOW level.

FIG. 9A is a circuit diagram of a multiplier and limiter circuit for apower converter circuit as shown in FIG. 7 and configured to operate incritical current mode.

In the critical current mode, the current mirror 98 is configured to setthe I_(VAL) signal to zero. In addition, the output of the comparator 92is multiplied by the I_(FF) signal at the multiplier 99.

FIG. 9B is a conceptual timing diagram illustrating the timing ofvarious voltage and current signals within a power converter circuit asshown in FIG. 7 and configured to operate in critical current mode. Asshown therein, when the current signal I_(CS) drops below zero a LOWvoltage is induced at the input of the inverter 112 and a high voltageis induced at the output of the inverter 112, which sets the PWM signalHIGH. When the current signal I_(CS) rises above the value of I_(PK), aHIGH voltage is induced at the input and output of the buffer 114, whichresets the PWM signal LOW.

In the hysteretic current mode, the resistor R7 is connected to groundwhich results in I_(FF)=0 and CrCM=LOW. Both I_(PK) and I_(VAL) arerelated to V_(COMP) by a fixed gain. The maximum I_(PK) is limited whichalso limits the boost inductor current.

In the critical current mode, R7 is connected from HYS-FF pin toV_(RECT). The current signal I_(PK) is normally scaled to the product ofV_(COMP) and I_(FF) (I_(VAL) is zero with switch S4 open). Both themaximum and minimum of I_(PK) are limited in the critical current mode.The maximum I_(PK) is limited to limit the boost inductor current. A lowlimit for the minimum I_(PK) sets the minimum peak inductor currentduring the zero-crossing of the AC input.

An integrated circuit controller as described herein regulates LEDcurrent from an AC input power. The integrated circuit modulates acascade switch in a boost or SEPIC converter powered from a rectified ACinput. The integrated circuit can be configured for hysteretic orcritical current mode, for example, by connection of a resistor toground or to the rectified input voltage. The integrated circuit mayinclude an integrated lower FET (part of the cascade switch), and mayprovide a low quiescent bias current, return current sensing, and/or lowvoltage reference and thresholds. Additionally the integrated circuitmay reduce the power dissipated with low bias current and/or low voltagereferences. The integrated circuit may further operate with increasedefficiency by employing an enhancement mode MOSFET as a high voltageswitch in a cascade switch configuration, and operating the high voltageswitch in saturated mode rather than linear mode.

FIGS. 10-16 are flowcharts illustrating operations of circuits/methodsaccording to some embodiments.

In particular, FIG. 10 illustrates a method of operating a voltageconversion circuit including a voltage boost circuit having a boostinductor according to some embodiments. The method includes receiving aninput voltage (block 202) and controlling operation of the voltage boostcircuit in response to a level of current in the boost inductor (block204).

FIG. 11 illustrates operations according to some embodiments. Referringto FIGS. 3, 5 and 11, the methods may include generating a first current(I_(CS)) that is representative of the current in the boost inductor(block 212), generating a current sense signal (i.e., the voltage acrossresistor R7) in response to the first current (block 214), comparing thecurrent sense signal to a threshold voltage defined by the ERROR signal(block 216), and changing a state of the pulse width modulation signalPWM in response to the comparison (block 218).

Referring to FIGS. 3 and 12, the methods may include generating a secondcurrent (I_(HYS)) (block 220), adding the first current (I_(CS)) to thesecond current to generate a combined current (I_(HYS)+I_(CS)) (block222), generating a combined voltage signal in response to the combinedcurrent (block 224), comparing the combined voltage signal to thethreshold voltage (block 226), and changing a state of the pulse widthmodulation signal in response to the comparison (block 228).

Referring to FIGS. 5 and 13, the methods may include generating acomparison signal COMP in response to a load current (block 230),generating a feedforward voltage signal in response to a feedforwardsignal (I_(FF)) that is representative of a level of a rectified inputvoltage signal (block 232), multiplying the comparison signal by thefeedforward signal to obtain an error signal ERROR (block 234),comparing the current sense signal to the error signal (block 236), andchanging a state of the pulse width modulation signal in response to thecomparison (block 238).

Referring to FIGS. 7 and 14, the methods may include generating acurrent signal (I_(CS)) that is representative of the current in theboost inductor (block 240) and changing a state of the pulse widthmodulation signal in response to the current signal falling below afirst reference current (I_(VAL)) or exceeding a second referencecurrent (I_(PK)) (block 242).

Referring to FIGS. 7, 8A and 15, the methods may include generating afirst current (I_(CS)) that is representative of the current in theboost inductor (block 250), generating a second current (I_(HYS)) havinga predetermined level (block 252), adding the first current to thesecond current to form a combined current (block 254), and changing astate of the pulse width modulation signal in response to the combinedcurrent falling below a reference current or in response to the firstcurrent exceeding the reference current (block 256).

Referring to FIGS. 7, 9A and 16, the methods may include generating afirst current (I_(CS)) that is representative of the current in theboost inductor (block 260), generating an error signal in response to aload current (block 262), generating first and second reference currents(I_(VAL), I_(PK)) in response to the error signal (block 264), andchanging a state of the pulse width modulation signal in response to thefirst current falling below the first reference current or in responseto the first current exceeding the second reference current (block 266).

Referring again to FIG. 7, circuits for performing the multiplication ofV_(COMP) by I_(FF) to obtain I_(PK) and I_(VAL) are known. For example,analog current signals can be converted to voltage signals using a senseresistor, and the resulting signals can be multiplied by usinglogarithmic amplifiers using the formula a*b=antilog(log(a)+log(b)).Analog signals can also be multiplied using transconductance amplifiers,such as Gilbert cells. However, each of these approaches may havedrawbacks in certain applications, such as power converters. Forexample, logarithmic amplifiers may present difficulties due to rangecompression, while transconductance amplifiers may have a very smalldynamic range, and may be susceptible to noise. Either of theseapproaches may be difficult to implement with a sufficient level ofaccuracy.

Some embodiments provide charge balance multipliers that can be used tomultiply an analog current by an analog voltage with a high level ofaccuracy. These multipliers may be used to multiply V_(COMP) by I_(FF)to obtain I_(PK) and I_(VAL) in power converters according to someembodiments. For example some embodiments may have an accuracy definedby the accuracy of a reference voltage, which can be very tightlycontrolled. Other embodiments may have an accuracy defined by the ratioof resistance of resistors, which can also be tightly controlled.

FIG. 17 illustrates a charge balance multiplier circuit 300 that may beused to implement the multiplier and limiter circuit 110 shown in FIG.7. Referring to FIG. 17, the charge balance amplifier circuit 300includes a feedforward conditioning circuit 305, a switched currentsource 310, a balance capacitor C_(BAL), a clamping diode V_(CLAMP), acontrol signal generating circuit 320 and an output current mirror 325including an output resistor R. The control signal generating circuit320 includes a ramping voltage generator 322 and a comparator 324. Thecontrol signal generating circuit 320 generates a control signal CTRLthat controls operation of the switched current source 310. The switchedcurrent source 310 may be implemented as a switched current mirror asshown in FIG. 17. However, it will be appreciated that other types ofcircuits may be used to provide the switched current source.

The feedforward conditioning circuit 305 includes a first current mirrorM1 and a second current mirror M2. The first current mirror M1 receivesa constant current signal I_(FF-Min) as an input. The feedforwardcurrent I_(FF) is input to the second current mirror M2. Outputs of boththe first and second current mirrors M1, M2 are coupled to the input ofthe switched current source 310. The first current mirror M1 ensuresthat a minimum level of feedforward current is drawn through theswitched current source 310. Accordingly, the input to the switchedcurrent source 310 can be expressed as max(I_(FF), I_(FF-Min)).

The output of the switched current source 310 is controlled by a controlsignal CTRL output by the control signal generating circuit 320, and inparticular generated by the comparator 324. When the control signal CTRLsignal is HIGH, the switched current source 310 outputs a current equalto the input current, which is equal to max(I_(FF), I_(FF)- _(Min)).When the control signal CTRL is LOW, the switched current source 310 isoff. The average charging current that is input to the balance capacitorC_(EA) may therefore be expressed as D*I_(CH), where I_(CH)=max(I_(FF),I_(FF)-_(min)) and D is the duty cycle of the control signal CTRL.

The amount of charge that is input to the balance capacitor C_(BAL) istherefore proportional to the duty cycle of the control signal CTRLoutput by the control signal generating circuit 320. The duty cycle ofthe control signal CTRL generated by the control signal generatingcircuit 320 is controlled by the charging and discharging cycle of aramping capacitor C_(ramp) in the control signal generating circuit 320and by the level of the input voltage V_(COMP).

The voltage at the output of the balance capacitor C_(BAL) is clamped bythe clamping diode V_(CLAMP), which limits the upper level of the I_(PK)current to I_(PK-HIGH). In the critical current mode, an extra thresholdvoltage V_(TH) is added by the transistor Q2, which is switched inresponse to the CrCM signal.

FIGS. 18A and 18B are diagrams that illustrate hypothetical curves forV_(RAMP), V_(COMP) and the current I_(CH) output by the switched currentsource 310.

Referring to FIGS. 17 and 18A, a constant current source I_(RAMP)charges the ramping capacitor C_(RAMP), causing the voltage level on theramping capacitor C_(RAMP) to rise in a linear fashion. When the voltageV_(RAMP) on the ramping capacitor exceeds V_(COMP), the output of thecomparator 324 goes HIGH. When the voltage V_(RAMP) exceeds the value ofV_(LIMIT), the output of the hysteretic comparator 323 transitions toHIGH, causing the transistor Q1 to turn on, which discharges the rampingcapacitor C_(RAMP). The hysteretic comparator 323 has a built-inhysteresis to allow the ramping capacitor C_(RAMP) to discharge to asuitably low level before being charged again.

The charge on the balance capacitor C_(BAL) is discharged through theoutput current mirror 325. Note that in the critical current mode, theI_(VAL) output is not used; accordingly it is switched out of thecurrent mirror 325 when the CrCM signal is HIGH. Because the charge thatis stored into the balance capacitor C_(BAL) must equal the charge thatis drawn from the capacitor, a charge balance equation may be written asfollows:(I _(CH) −I _(PK))DT _(S) =I _(PK)(1−D)T _(S)  (1)

where Ts is the switching period.

From equation (1), it is apparent that the duty cycle D can be expressedas a function of I_(PK) and I_(CH) as follows:D=I _(PK) /I _(CH)  (2)

However, the duty cycle can also be expressed as a function of thevoltages V_(COMP) and V_(LIMIT) as follows:D=V _(COMP) /V _(LIMIT)  (3)

Combining equations (2) and (3) yields an expression for I_(PK) in termsof I_(FF) and V_(COMP) as follows:I _(PK)=(V _(COMP) *I _(CH))N _(LIMIT)  (4)

Because the charging current I_(CH) is simply the feedforward currentI_(FF) with a floor of I_(FF-Min), I_(PK) is proportional to the productof the feedforward current I_(FF) and the voltage V_(COMP).

Referring to FIG. 18B, as the voltage V_(COMP) increases, the duty cycleD of the charging current I_(CH) increases. When V_(COMP) reachesV_(LIMIT), the duty cycle D becomes unity, placing an upper limit on theamount of charge that can be stored into the balance capacitor C_(BAL),and effectively placing an upper limit on I_(PK). This is illustratedgraphically in FIG. 19, which shows that the duty cycle D increaseslinearly as V_(COMP) increases up to the value of V_(LIMIT).Accordingly, in some embodiments, the clamping diode D5 shown in FIG. 7may not be needed.

In the hysteretic mode, the feedforward current I_(FF) is zero, so theminimum feedforward current I_(FF-Min) is drawn through the PWMcontrolled current mirror 310. Moreover, I_(VAL) is also generated bythe output current mirror 325 in the hysteretic mode, and is equal tothe I_(PK) current.

A charge balance multiplier circuit 400 according to further embodimentsis illustrated in FIG. 20. The charge balance multiplier circuit 400includes a current mirror 405 that generates a charging signal I_(CH)that charges a balance capacitor C_(BAL) at node N. The charging signalI_(CH) is equal to max(I_(FF), I_(FF-Min)), which may be generated, forexample, using a feedforward conditioning circuit 305 as illustrated inFIG. 17. The balance capacitor C_(BAL) is discharged by a full scalecurrent reference I_(FS) through a switched current source 410.

The balance capacitor C_(BAL) is coupled at node N to an input of ahysteretic inverter 420. The signal input to the hysteretic inverter 420is equal to the voltage on the balance capacitor C_(BAL), which isdetermined by a full scale current reference I_(FS) and the chargingcurrent I_(CH).

The balance capacitor C_(BAL) is charged by the charging current I_(CH)when the voltage on the balance capacitor C_(BAL) is lower than thelevel needed to force the output of the hysteretic inverter 420 low. Inthat case, the output of the hysteretic inverter 420 is high, whichturns on the transistor switch Q3 and turns off the switched currentsource 410. When the switched current source 410 is turned off, the fullscale reference current I_(FS) is not drawn from node N by the switchedcurrent source 410.

When the charging current I_(CH) charges the balance capacitor to avoltage level that is sufficient to force the output of the hystereticinverter 420 low, the transistor switch Q3 turns off, which activatesthe switched current source 410 and causes a current equal to the fullscale reference current I_(FS) to be drawn from node N. This results ina discharge of current from the balance capacitor C_(BAL). Because thecharging current I_(CH) continues to flow, the discharge current fromthe balance capacitor C_(BAL) is given as I_(FS)−I_(CH).

The output of the hysteretic inverter 420 is provided as an input to aCMOS inverter 43Q. Thus, the signal input to the CMOS inverter 430 has aduty cycle of 1−D.

As described above, the discharge current from the balance capacitorC_(BAL) is switched by the transistor switch Q3 at the 1−D duty cycle inresponse to the output of the hysteretic inverter 420. Since the chargeflowing into the balance capacitor C_(BAL) when it is being charged isequal to the charge flowing out of the balance capacitor C_(BAL) when itis being discharged, a charge balance equation may be written asfollows:(I _(FS) −I _(CH))DTs=I _(CH)(1−D)Ts  (5)

From equation (5), it is possible to express the duty cycle D in termsof I_(CH) and I_(FS) as follows:D=I _(CH) /I _(FS)  (6)

The CMOS inverter 430 generates an output voltage that alternatesbetween ground and V_(COMP). The CMOS inverter 430 therefore generates apulse train that has a duty cycle of D that is proportional to I_(CH)and that has an amplitude between ground and V_(COMP).

The signal output by the CMOS inverter 430 is filtered by an RC filter435, which averages the pulse train to form a voltage having a level of(I_(CH)*V_(COMP)/I_(FS)). The signal is then amplified by an amplifier440 having a gain K to generate the current I_(PK) through currentmirrors M3, M4. The voltage signal is clamped at a level of V_(CLAMP),which places an upper limit I_(PK-HIGH) on I_(PK). That is:V _(CLAMP) =I _(PK-HIGH) *R  (7)whereR=1/(I _(FS) *K)  (8)

The gain K of the amplifier 440 is set so that I_(PK)=I_(CH)*V_(COMP)*K.

In hysteretic mode, I_(FF) is equal to zero, so the switch S1 remainsclosed to shut off the current reference I_(FS). The I_(FF-MIN)reference signal saturates the hysteretic comparator 420 HIGH, and theoutput of the CMOS inverter 430 stays at V_(COMP). I_(PK) and I_(VAL)are therefore set at V_(COMP)/R.

The circuits/methods described herein for performing themultiplying/limiting function of the multiplier and limiter 110 of FIG.7 may provide highly linear first quadrant analog multiplication of acurrent signal and a voltage signal. These circuits/methods may providea very accurate solution for the multiplying/limiting function, whichmay be important for the operation of a power conversion circuit asdescribed herein.

Although the circuits illustrated in FIG. 17 and are shown as beingimplemented with MOSFET transistor switches, it will be appreciated thatother types of transistor switches, such bipolar junction transistor(BJT) switches could be used in some embodiments.

Many different embodiments have been disclosed herein, in connectionwith the above description and the drawings. It will be understood thatit would be unduly repetitious and obfuscating to literally describe andillustrate every combination and subcombination of these embodiments.Accordingly, all embodiments can be combined in any way and/orcombination, and the present specification, including the drawings,shall be construed to constitute a complete written description of allcombinations and subcombinations of the embodiments described herein,and of the manner and process of making and using them, and shallsupport claims to any such combination or subcombination.

In the drawings and specification, there have been disclosed typicalembodiments and, although specific terms are employed, they are used ina generic and descriptive sense only and not for purposes of limitation,the scope of the inventive concepts being set forth in the followingclaims.

What is claimed is:
 1. A circuit for generating an output current,comprising: a control signal generating circuit that is configured togenerate a control signal, wherein the control signal is a function of alevel of an analog input voltage signal and wherein an average level ofthe output current is a function of a level of a varying analog inputcurrent signal and the level of the analog input voltage signal; and aswitched current source that is controlled by the control signal andconfigured to generate the output current with a duty cyclecorresponding to the control signal and having a peak level equal to thelevel of the varying analog input current signal, wherein the duty cycleof the control signal is proportional to the level of the analog inputvoltage signal.
 2. The circuit of claim 1, further comprising: a balancecapacitor coupled to the switched current source; wherein a level of theoutput current is proportional to a product of the level of the varyinganalog input current signal and the level of the analog input voltagesignal.
 3. The circuit of claim 2, further comprising: an output currentmirror coupled to the balance capacitor; wherein the switched currentsource receives the varying analog input current signal as an input; andwherein the balance capacitor is charged by the switched current withthe duty cycle corresponding to the control signal and formed from thevarying analog input current signal output by the switched currentsource and is discharged through the output current mirror.
 4. Thecircuit of claim 3, further comprising a resistor between the balancecapacitor and the output current mirror.
 5. The circuit of claim 2,wherein the control signal generating circuit comprises: a comparatorconfigured to receive the analog input voltage signal and a ramp voltageand to generate the control signal in response to the analog inputvoltage signal and the ramp voltage; and a ramp voltage generatingcircuit coupled to the comparator and configured to generate the rampvoltage.
 6. The circuit of claim 5, wherein the ramp voltage generatingcircuit comprises: a current source configured to generate a rampingcurrent; a ramping capacitor coupled to the current source andconfigured to be charged by the ramping current; a transistor switchconfigured to discharge a ramping voltage from the ramping capacitor inresponse to a discharge signal; and a hysteretic comparator configuredto compare the ramping voltage with a reference voltage and to generatethe discharge signal in response to a comparison of the ramping voltagewith the reference voltage.
 7. The circuit of claim 5, wherein theswitched current source comprises: a first transistor having a gateterminal and a drain terminal; a second transistor having a gateterminal coupled to the gate terminal of the first transistor and adrain terminal, coupled to the drain terminal of the first transistor;and a switch transistor coupled between the drain terminals of the firstand second transistors and the gate terminals of the first and secondtransistors, wherein the switch transistor has a gate terminal coupledto an output of the comparator and is configured to receive the controlsignal.
 8. The circuit of claim 2, farther comprising an input currentconditioning circuit comprising a first current mirror and a secondcurrent mirror coupled to the first current mirror, wherein the firstcurrent mirror is configured to supply an input current signal as thevarying analog input current signal when the varying analog inputcurrent signal is above a threshold level, and wherein the secondcurrent mirror is configured to supply a reference current signal as thevarying analog input current signal when the varying analog inputcurrent signal is below the threshold level.
 9. The circuit of claim 2,wherein the circuit for generating the output current is configured tobe switched between a first mode in which the input current signal isnonzero and a second mode in which the varying analog input currentsignal is zero.
 10. The circuit of claim 2, further comprising aclamping diode coupled to the balance capacitor.
 11. The circuit ofclaim 2, wherein the output current is given as:I _(PK)=(V _(COMP) *I _(CH))/V _(LIMIT) where I_(PK) is the outputcurrent V_(COMP) is the analog input voltage signal, I_(CH) is theswitched current with the duty cycle corresponding to the control signaland formed from the varying analog input current signal, and V_(LIMIT)is a reference voltage.
 12. The circuit of claim 11, wherein a chargethat is stored in the balance capacitor is given as (I_(CH)−I_(PK))DTs,where D is the duty cycle of the control signal and Ts is a period ofthe control signal, and wherein a charge that is discharged from thebalance capacitor is given as I_(PK)(1−D)Ts.
 13. The circuit of claim 2,wherein the switched current source receives a reference current as aninput, and wherein the balance capacitor is charged by the varyinganalog input current signal and is discharged by the switched currentsource.
 14. The circuit of claim 13, further comprising: a hystereticinverter having an input coupled to the balance capacitor; and a switchcoupled to an output of the hysteretic inverter and configured tocontrol the switched current source.
 15. The circuit of claim 14,further comprising: an output inverter having an input coupled to theoutput of the hysteretic inverter and configured to generate an outputsignal having an amplitude that is proportional to the analog inputvoltage signal.
 16. The circuit of claim 15, further comprising: afilter configured to filter the output signal of the output inverter;and an amplifier configured to amplify the filtered output signal of theoutput inverter.
 17. The circuit of claim 13, wherein the output currentsignal is given by:I _(PK) =I _(CH) *V _(COMP) *K wherein I_(PK) is the output current,I_(CH) is the switched current with the duty cycle corresponding to thecontrol signal and formed from the varying analog input current signal,V_(COMP) is the analog input voltage signal, and K is a constant. 18.The circuit of claim 17, wherein a charge that is stored in the balancecapacitor is given as I_(CH)(1−D)Ts, where D is a duty cycle of thecontrol signal and Ts is a period of the control signal, and wherein acharge that is discharged from the balance capacitor is given as(I_(FS)−I_(CH))DTs, wherein I_(FS) is the reference current.
 19. Acircuit for generating an output current, comprising: a control signalgenerating circuit that generates a control signal in response to ananalog input voltage; a switched current source that generates theoutput current in response to the control signal and a variable analoginput current signal; and a balance capacitor coupled to an output ofthe switched current source; wherein a duty cycle of the control signalis proportional to a level of the analog input voltage signal; andwherein a level of the output current is proportional to a product ofthe variable analog input current signal and the analog input voltage;an output current mirror coupled to the balance capacitor; wherein thebalance capacitor is charged by a current output by the switched currentsource and is discharged through the output current mirror.
 20. Thecircuit of claim 19, wherein the control signal generating circuitcomprises: a comparator configured to receive the analog input voltageand a ramp voltage and to generate the control signal in response to theanalog input voltage and the ramp voltage; and a ramp voltage generatingcircuit coupled to the comparator and configured to generate the rampvoltage.
 21. A circuit for generating an output current, comprising: acontrol signal generating circuit including a comparator configured toreceive an analog input voltage signal and a ramp voltage and togenerate a control signal in response to the analog input voltage signaland the ramp voltage; and a switched current source that generates theoutput current in response to the control signal and a varying analoginput current signal, wherein the current output by the switched currentsource is formed from the varying analog input current signal and has aduty cycle that corresponds to a duty cycle of the control signal;wherein the duty cycle of the control signal is proportional to a levelof the analog input voltage signal; and wherein a level of the outputcurrent is proportional to a product of the varying analog input currentsignal and the analog input voltage signal; wherein the circuit furthercomprises: a balance capacitor coupled to an output of the switchedcurrent source; and an output current mirror coupled to the balancecapacitor; wherein the balance capacitor is charged by the currentoutput by the switched current source and is discharged through theoutput current mirror.